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PDF LTC3605 Data sheet ( Hoja de datos )
| Número de pieza | LTC3605 | |
| Descripción | 5A Synchronous Step-Down Regulator | |
| Fabricantes | Linear Dimensions Semiconductor | |
| Logotipo |  |
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FEATURES
n High Efficiency: Up to 96%
n 5A Output Current
n 4V to 15V VIN Range
n Integrated Power N-Channel MOSFETs
(70mΩ Top and 35mΩ Bottom)
n Adjustable Frequency 800kHz to 4MHz
n PolyPhase® Operation (Up to 12 Phases)
n Output Tracking
n 0.6V ±1% Reference Accuracy
n Current Mode Operation for Excellent Line and Load
Transient Response
n Shutdown Mode Draws Less than 15μA Supply
Current
n Available in 24-Pin (4mm × 4mm) QFN Package
APPLICATIONS
n Point of Load Power Supply
n Portable Instruments
n Distributed Power Systems
n Battery-Powered Equipment
LTC3605
15V, 5A Synchronous
Step-Down Regulator
DESCRIPTION
The LTC®3605 is a high efficiency, monolithic synchronous
buck regulator using a phase lockable controlled on-time
constant frequency, current mode architecture. PolyPhase
operation allows multiple LTC3605 regulators to run out of
phase while using minimal input and output capacitance.
The operating supply voltage range is from 15V down to
4V, making it suitable for dual lithium-ion battery inputs
as well as point of load power supply applications from
a 12V or 5V rail.
The operating frequency is programmable from 800kHz to
4MHz with an external resistor. The high frequency capabil-
ity allows the use of small surface mount inductors. For
switching noise sensitive applications, it can be externally
synchronized from 800kHz to 4MHz. The PHMODE pin
allows user control of the phase of the outgoing clock
signal. The unique constant frequency/controlled on-time
architecture is ideal for high step-down ratio applications
that are operating at high frequency while demanding
fast transient response. Two internal phase-lock loops
synchronize the internal oscillator to the external clock
and also servos the regulator on-time to lock on to either
the internal clock or the external clock if it’s present.
L, LT, LTC, LTM and Polyphase are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners. Protected by U.S. Patents,
including 5481178, 5847554, 6580258, 6304066, 6476589, 6774611.
TYPICAL APPLICATION
VIN
4V TO 15V
High Efficiency 1MHz, 5A Step-Down Regulator
22μF
s2
VIN
PVIN
SVIN
CLKOUT INTVCC
CLKIN
PGOOD BOOST
LTC3605
SW
VON
RUN FB
ITH
RT
162k
2.2μF
0.1μF L
11.5k
16k
220pF
3605 TA01a
2.55k
VOUT
3.3V
47μF
s2
PGND
Efficiency and Power Loss
100
VIN = 12V
90 f = 1MHz
VOUT = 3.3V
10
80
70
VOUT = 1.2V
60
50
1
40
30
VOUT = 3.3V VOUT = 1.2V
0
20
10
0 0.1
10
100
1000
10000
OUTPUT CURRENT (mA)
3605 TA01b
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LTC3605
TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C unless otherwise specified.
Switch Leakage vs VIN
350
300
250 BOTTOM SWITCH
200
150
100
50
TOP SWITCH
0
0 4 8 12 16 20
VIN (V)
3605 G10
Efficiency vs VIN
98
VOUT = 3.3V
96
ILOAD = 1A
94
92
90
88 ILOAD = 5A
86
84
0 3 6 9 12 15
VIN (V)
3605 G11
Frequency vs VON Voltage
1.6
VIN = 14V
1.4 RT = 162k
1.2
1.0
0.8
0.6
0.4
0.2
0
0 2 4 6 8 10 12 14
VON (V)
3605 G12
Current Limit Foldback
120
100
80
60
40
20
0
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7
VFB (V)
3605 G13
INTVCC Load Regulation
101
100 TA = 25°C
99
98
97
96
95
94
93
92
91
90
0 20 40 60 80 100 120 140
INTVCC OUTPUT CURRENT (mA)
3605 G14
RUN Pin Threshold vs Temperature
1.30
1.25
1.20
1.15
1.10
1.05
1.00
–40 –15
10 35 60 85
TEMPERATURE (°C)
110
3605 G15
DCM Operation
CLKOUT
2V/DIV
VSW
5V/DIV
IL
2A/DIV
VIN = 12V
VOUT = 1.2V
MODE = 0
IOUT = 0
L1 = 0.33μH
400ns/DIV
3605 G16
CCM Operation
CLKOUT
2V/DIV
VSW
5V/DIV
IL
2A/DIV
VIN = 12V
VOUT = 1.2V
MODE = 3.3V
PHMODE = 3.3V
IOUT = 0
L1 = 0.33μH
400ns/DIV
3605 G17
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LTC3605
OPERATION
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. However, care must
be taken when these capacitors are used at the input and
output. When a ceramic capacitor is used at the input
and the power is supplied by a wall adapter through long
wires, a load step at the output can induce ringing at the
VIN input. At best, this ringing can couple to the output and
be mistaken as loop instability. At worst, a sudden inrush
of current through the long wires can potentially cause a
voltage spike at VIN large enough to damage the part.
When choosing the input and output ceramic capacitors,
choose the X5R and X7R dielectric formulations. These
dielectrics have the best temperature and voltage charac-
teristics of all the ceramics for a given value and size.
Since the ESR of a ceramic capacitor is so low, the input
and output capacitor must instead fulfill a charge storage
requirement. During a load step, the output capacitor must
instantaneously supply the current to support the load
until the feedback loop raises the switch current enough
to support the load. The time required for the feedback
loop to respond is dependent on the compensation and the
output capacitor size. Typically, 3 to 4 cycles are required
to respond to a load step, but only in the first cycle does
the output drop linearly. The output droop, VDROOP, is
usually about 2 to 3 times the linear drop of the first cycle.
Thus, a good place to start with the output capacitor value
is approximately:
COUT
≈ 2.5
fO
ΔIOUT
• VDROOP
More capacitance may be required depending on the duty
cycle and load step requirements.
In most applications, the input capacitor is merely required
to supply high frequency bypassing, since the impedance to
the supply is very low. A 22μF ceramic capacitor is usually
enough for these conditions. Place this input capacitor as
close to the PVIN pins as possible.
Inductor Selection
Given the desired input and output voltages, the induc-
tor value and operating frequency determine the ripple
current:
ΔIL
=
VOUT
f •L
⎛
⎜ 1–
⎝
VOUT
VIN(MAX )
⎞
⎟
⎠
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors and output voltage
ripple. Highest efficiency operation is obtained at low
frequency with small ripple current. However, achieving
this requires a large inductor. There is a trade-off between
component size, efficiency and operating frequency.
A reasonable starting point is to choose a ripple current
that is about 50% of IOUT(MAX). This is especially impor-
tant at low VOUT operation where VOUT is 1.8V or below.
Care must be given to choose an inductance value that
will generate a big enough current ripple (40% to 50%)
so that the chip’s valley current comparator has enough
signal-to-noise ratio to force constant switching frequency.
Meanwhile, also note that the largest ripple current occurs
at the highest VIN. To guarantee that ripple current does
not exceed a specified maximum, the inductance should
be chosen according to:
L
=
f
•
VOUT
ΔIL(MAX )
•
⎛
⎜ 1–
⎝
VOUT
VIN(MAX )
⎞
⎟
⎠
Once the value for L is known, the type of inductor must
be selected. Actual core loss is independent of core size
for a fixed inductor value, but is very dependent on the
inductance selected. As the inductance or frequency in-
creases, core losses decrease. Unfortunately, increased
inductance requires more turns of wire and therefore
copper losses will increase.
Ferrite designs have very low core losses and are pre-
ferred at high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard”, which means that
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| Páginas | Total 20 Páginas | |
| PDF Descargar | [ Datasheet LTC3605.PDF ] | |
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